FIG. 1 is a schematic block diagram of a wireless transmitter. The transmitter comprises an initial data processing stage 2 arranged to generate data on an in-phase (I) branch and a quadrature branch (Q). In operation, the data processing stage 2 begins by generating digital data for one or more channels to be modulated for transmission via the I branch and one or more channels to be modulated for transmission via the Q branch.
The data on each channel is initially generated in binary form. As a first step of the modulation, the data processing stage 2 then maps each of a sequence of binary data portions from that channel onto an actual value. For example, in a binary phase-shift keying (BPSK) scheme each logical 0 in the sequence is mapped onto an actual value −1 and each logical 1 is mapped onto an actual value +1. Or as another example, in a 4-QAM scheme each two-bit portion of binary data 00, 01, 10 or 11 in the sequence is mapped onto one of a set of four actual values such as 0.4472, 1.3416, −0.4472 and −1.3416 respectively.
Each value is then multiplied by a higher-rate spreading code comprising multiple higher-rate “chips” for each un-spread value, with a different respective spreading code being used for each channel according to a code division multiple access (CDMA) scheme. As will be familiar to a person skilled in the art, since each channel is spread by a different orthogonal spreading code, this allows different channels from the same transmitter and/or other transmitters to be multiplexed over the same frequency at the same time in the same physical space yet still be separately identified at the receiver(s). Also, since the spreading code comprises multiple chips for each un-spread value then the sample rate is increased to a higher rate referred to as the “chipping” rate, equal to 3.84Msps in current WCDMA standards.
Each channel is then also weighted by a respective weighting factor. Finally, if there are a plurality of channels to be modulated via the I branch then the data processing stage 2 multiplexes the spread, weighted data from those channels into a combined I component which it outputs onto the I branch. If there are a plurality of channels to be modulated via the Q branch then the spread, weighted data from those channels is multiplexed into a combined Q component which is output onto the Q branch. (Or alternatively the I and/or Q component could result from only a single I or Q channel respectively.
Details of the above mapping, spreading and weighting steps are provided, for example, in 3GPP TS 25.213, Technical Specification Group Radio Access Network, “Spreading and Modulation (FDD)”.
The transmitter further comprises a pulse-shape filter 4 comprising a constituent I-branch pulse-shape filter 4i and Q-branch pulse-shape filter 4q. The output from the data processing stage 2 on the in-phase (I) branch is coupled to the input of the I-branch pulse-shape filter 4i, and the output from the data processing stage 2 on the quadrature branch (Q) is coupled to the input of a Q-branch pulse-shape filter 4q. A pulse-shape filter 4 may also be referred to as a shaping filter or sometimes a channel filter. Each constituent pulse-shape filter 4i, 4q typically operates in the time domain. Each of the constituent pulse-shape filters 4i, 4q may be a Root Raised Cosine (RRC) filter parameterized by a roll-off factor α of −0.22 according to current WCDMA standards. The impulse response H(t) of such a filter is shown schematically in the sketch of FIG. 2. Thus the I-branch pulse-shape filter 4i filters the I component generated by the data processing stage 2 so as to output a filtered version of the I component filtered according to H(t). Similarly the Q-branch pulse-shape filter 4q filters the Q component generated by the data processing stage 2 so as to output a filtered version of the Q component filtered according to H(t). Thus for each data sample input into the pulse-shape filter 4i or 4q representing the unshaped I or Q component at some point in time, the pulse-shape filter 4 spreads the effect of that data over time according to a pulse shape defined by the function H(t), with the effects from the samples from different times being superimposed with one another. The operation of such a time-domain filter 4 will be familiar to a person skilled in the art.
The output of the I-branch filter 4i is coupled to the input of a cosine modulation block 6i where the filtered version of the I component is modulated by a cosine carrier so as to output a modulated version of the I component on the I branch. The output of the Q-branch filter 4q is coupled to the input of a sine wave modulation block 6q where the filtered version of the Q component is modulated by a sine wave carrier so as to output a modulated version of the Q component on the Q branch. Because the Q component is modulated by a carrier that is 90° (π/2 radians) out of phase with the carrier of the I component, then the I and Q components together may be considered to form a complex signal which may be represented mathematically by an expression of the form I+jQ.
The outputs of the cosine and sine modulation blocks 6i and 6q are coupled to respective inputs of an adder 8, where the shaped, modulated I and Q components from the two respective branches are summed so as to generate a combined output for transmission. Finally this combined output is coupled to a power amplifier (PA) 12 via an upconversion mixer stage and analog-to-digital converter (not shown), such that the combined output signal is mixed up to radio frequency, converted to analog and then amplified for wireless transmission via an antenna 14. These techniques are typically used for transmission over a wireless cellular network.
Recent cellular radio communications standards such as LTE, WiMAX and the HSPA+ and HSUPA parts of WCDMA specify uplink modulation schemes that can result in a relatively high Peak-to-Average-Ratio (PAR) of power compared with older cellular communications standards such as GSM/EDGE and Release-99 WCDMA.
At the same time, very stringent requirements exist for limiting the Adjacent Channel Leakage Ratio (ACLR)—that is, the level of unwanted emissions in the adjacent radio channel. Also, the signal must be distorted as little as possible; this requirement is expressed as a limit on the Error Vector Magnitude (EVM) of the transmitted uplink signal.
Generally speaking, increased non-linearity in the radio frequency power amplifier (PA) leads to increased adjacent channel leakage and distortion.
Achieving PA linearity while maximizing PA power efficiency becomes easier when the PAR is lower (and vice-versa).
Thus if the uplink signal could be somehow pre-distorted so as to lower the PAR of the signal amplified by the PA 12 without degrading the EVM unacceptably, then it would be possible to make the PA more power efficient. This is particularly important in hand-held and portable equipment (e.g. smartphones, netbooks, etc.) where a significant amount of battery power is consumed by the PA and any reduction would be beneficial, both in terms of extending the time for which the equipment can be used before re-charging the battery and in terms of limiting the amount of heat generated by the PA itself.
Some pre-processing schemes for limiting PAR have been proposed already. One such scheme is disclosed in “Algorithm for Peak to Average Power Ratio Reduction operating at Symbol Rate”, by Stefano Marsili, Infineon Technology, Austria AG, Villach, Austria. IEEE ISCAS 2005. (Hereinafter referred to as Marsili.) However, this scheme has a high computational complexity and hence incurs a high processing burden in terms of processing cycles. It was also designed for the downlink signal which historically has always had a much higher PAR than the uplink signal, and it cannot be assumed that such a scheme would be applicable on the uplink whilst still meeting strict ACLM and EVM requirements.